Digital induction logging system including means for generating a plurality of transmitter frequencies

ABSTRACT

A digital induction resistivity logging system is disclosed for digitally measuring phase components in a receiver signal generated in response to eddy currents flowing in the earth&#39;s sub-surface formations. The induction logging system includes a central processing unit (CPU) located at the surface for receiving digital signals from a digital induction logging tool located downhole, and for analyzing the received data. The CPU supplies command and control signals to the induction logging tool to specify operating modes and parameters for obtaining the digital signals. The system further includes a digital telemetry system associated with a wireline cable for transmitting and receiving data between the induction tool and the CPU. The digital induction tool includes a digital sinewave generator for generating a highly phase stable, low distortion transmitter signal whose frequency is selectable from at least two of transmitter frequencies. Selection of the transmitter frequency may be based on optimizing the measurement of a characteristic of the formations being encountered by the tool. Automatic phase compensation is included to dynamically compensate for both static and dynamic temperature dependent phase errors due to circuits of the tool involved in the component measurements. A floating point analog-to-digital convertor capable of responding to the wide dynamic range in the detected phase component signals is provided to convert the phase detector output into digital signals for use by the CPU.

TABLE OF CONTENTS

CROSS REFERENCE TO RELATED APPLICATIONS

BACKGROUND OF THE INVENTION

SUMMARY OF THE INVENTION

BRIEF DESCRIPTION OF THE DRAWINGS

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT OF THE INVENTION

The Digital Induction Logging System

The Digital Induction Logging Tool

The Controller

The Waveform Generator

The Phase Shift Error Compensation Circuits

The Autophase Unit

The Autocalibration Circuits

The Floating Point Analog-to-Digital Convertor

Summary of Operation

THE CLAIMS

CROSS REFERENCE TO RELATED APPLICATIONS

This application relates to the following applications which are filedconcurrently herewith: U.S. patent application No. 271,278 and entitled"A Digital Induction Logging System Including Means for Measuring PhaseQuadrature Components in a Phase Sensitive Detector; " U.S. patentapplication No. 271,279 and entitled "A Digital Induction Logging ToolIncluding a Floating-Point A/D;" U.S. patent application No. 271,280 andentitled "A Digital Induction Logging Tool Including Means forCompensating for Phase Shift Errors;" U.S. patent application No.271,367 and entitled "Digital Induction Tool;" and U.S. patentapplication No. 271,277 and entitled "An Induction Logging SystemIncluding Autocalibration." Each of the related applications areassigned to the same Assignee as the present application.

BACKGROUND OF THE INVENTION

This invention relates to electrical induction logging systems fordetermining the nature and characteristics of the various sub-surfaceformations penetrated by a borehole drilled into the earth. Moreparticularly, this invention relates to a digital induction loggingsystem for obtaining digital samples of signals characteristic of theresistivity of the formations.

It is important to the oil and gas industry to know the nature andcharacteristics of the various sub-surface formations penetrated by aborehole since the mere drilling of a borehole usually does not providesufficient information concerning the existence, depth location,quantity, etc., of oil and gas trapped in the formations. Variouselectrical techniques have been employed in the past to determine thisinformation about the formations. One such technique commonly used isinduction logging. Induction logging measures the resistivity (or itsinverse, conductivity) of the formation by first inducing eddy currentsto flow in the formations in response to an AC transmitter signal, andthen measuring a phase component signal in a receiver signal generatedby the presence of the eddy currents. Variations in the magnitude of theeddy currents in response to variations in formation conductivity arereflected as variations in the receiver signal. Thus, in general, themagnitude of a phase component of the receiver signal, that componentin-phase with the transmitter signal, is indicative of the conductivityof the formation.

In theory, the electrical resistivity of the formation should berelatively high when the formation contains a high percentage ofhydrocarbons because hydrocarbons are relatively poor conductors ofelectricity. Where hydrocarbons are not present in the formations andthe formations contain salt water, the electrical resistivity of theformation should be relatively low. Formation water, which typically issalty, is a relatively good conductor of electricity. Inductionresistivity logging tools thus obtain information about the formationswhich can be interpreted to indicate the presence or absence of thesehydrocarbons.

U.S. Pat. Nos. 3,340,464, 3,147,429, 3,179,879 and 3,056,917 areillustrative of typical prior-art well logging tools which utilize thebasic principles of induction logging. In each of the tools disclosed inthese patents, a signal generator operates to produce an AC transmittersignal which is applied to a transmitter coil. The current in thetransmitter coil induces a magnetic field in the formations. Thismagnetic field, in turn, causes eddy currents to flow in the formations.Because of the presence of these formation currents, a magnetic field iscoupled into a receiver coil R thereby generating a receiver signal.(Logging tools having "a receiver coil" and "a transmitter coil" eachcomprised of several coils arranged in a predetermined fashion to obtaina desired response are commonly used.) The receiver signal is thenamplified and applied to one or more phase sensitive detectors (PSDs).Each PSD detects a phase component signal having the same phase as aphase reference signal which is also applied to the detector. The phasereference signal has a predetermined phase relationship to the currentin the transmitter coil(s). The output of the PSD(s) may be furtherprocessed downhole, or may be sent uphole to surface equipment forprocessing or display to an operating engineer.

Heretofore, prior-art induction resistivity logging tools have beenprimarily analog in design, with some digital circuits used to performsome functions, e.g., see the digital flip-flops of U.S. Pat. No.3,340,464. Because of the analog nature of prior-art designs and forother reasons, these prior-art tools have limitations which prevent themfrom meeting a growing need for more precise, accurate and error freemeasurements of phase component signals in the receiver signal.

A quantitative determination of the conductivity of the formations isbased in large part on the value obtained for the phase component signalthat is inphase with the transmitter current in the transmitter coil.This component signal is referred to as the real or "R" phase component.Measurement of a phase component signal which has a phase orthogonal to(or in other words, in quadrature to) the transmitter current issometimes obtained. This component signal is referred to as the "X"phase component signal.

Measurement of both the R and X phase component signals of the receiversignal is known. U.S. Pat. Nos. 3,147,429 and 3,179,879 both discloseinduction logging tools which detect phase quadrature components (V_(r)and V_(x) ') of the receiver signal from the receiver coil. The toolsdisclosed in these patents show the output from a receiver amplifierbeing applied to identical PSD circuits, one for detecting the Rcomponent signal and the other for detecting the X component signal.Appropriate phase shifting components are provided for generating thephase quadrature phase reference signals required by the PSDs in orderto resolve the phase component signals.

The need for higher precision and accuracy in the resolution of thesephase component signals is a natural consequence of the need to knowmore about formation characteristics that can be extracted from thesignals representative of these characteristics. But, to obtain accuratemeasurements, the inaccuracies present in the measurements obtained bythe prior-art tools must be eliminated. A principal source ofinaccuracies in the measurement of the R and X component signals presentin prior-art logging tools results from phase shifts in the signals ofthe tool. These phase shifts result in a departure from thein-phase/quadrature phase relationship between the transmitter signal,the receiver signal and the phase reference signals, all of which areused in resolving the received signal into the quadrature componentsignals R and X.

Two principle sources of phase shift errors are present in inductionlogging tools--static phase shift errors and dynamic (temperaturedependent) phase shift errors. Static phase shift errors are those phaseshifts which occur when the tool is operating at a steady statetemperature condition. These phase shift errors are introduced into thedetected phase component signal by certain electrical circuits in thetool, i.e., the transmitter coil system, the receiver coil system, theamplifier used to condition the receiver signal and the PSD itself. Thedynamic phase shift errors occur as a result of such influences astemperature drift in these same circuits, all of which are involved inthe generation of the formation currents and in the detection of thephase components in the receiver signal. Unpredictable phase shifts mayalso be introduced by component variations that are an unavoidableconsequence of the manufacturing process. High precision resolution ofthe component signals requires that these phase shift errors beautomatically and periodically eliminated from the measurements duringthe logging operation. This is especially true since the temperatureenvironment in which the induction tool is operated will vary over awide range with the depth in the borehole.

The dynamic compensation for phase shift errors due to temperature driftin the circuits of an induction logging tool has been attempted in theprior art. U.S. Pat. No. 3,340,464 discloses a circuit for automaticallyadjusting for varying phase shifts due to temperature drift in thetool's circuits by deriving a test signal from the current in thetransmitter coil; substituting this test signal for the normal receivercoil output signal; generating a quadrature reference signal to the PSDto detect a phase component (X) in the receiver signal; and, phaseshifting the reference signal as a function of the magnitude of thedetected phase component signal in a direction to minimize that signal.This disclosed phase error compensation circuit and method does notattempt to segregate the relatively fixed or constant phase errors ofthe tool from the temperature dependent phase errors which vary withtime during logging and resulting from component drift in the circuits.Rather, the tool of U.S. Pat. No. 3,340,464 attempts to compensate forany and all phase shifts regardless of their source which have occurredsince the last phase compensation.

As a result, the phase compensation circuit of U.S. Pat. No. 3,340,464must compensate for the phase angle error over a greater range of anglesthan would be required if the static and temperature dependent phaseshift errors were separately compensated. A large range in phase anglecompensation results in less sensitivity to small phase shift errors.This loss of sensitivity allows uncompensated phase shift errors toappear in the detected phase component signal. These errors prohibit thehigh precision and accuracy in the measurements.

Those prior-art tools, such as those disclosed in U.S. Pat. Nos.3,147,429 and 3,179,879, which measure both R and X require two PSDs,one for measuring R and one for measuring X. This dual arrangement ofdetecting circuits in an induction tool implies that the static andtemperature dependent phase shift errors for each of the two PSD's willnot be the same, i.e., the circuits will not respond identically to agiven temperature change even if they could be made to have the samephase shift at a given temperature. Because of this difference,different phase shift errors will be present in the R and Xmeasurements. Even with phase shift compensation techniques, such asthat disclosed in U.S. Pat. No. 3,340,464, applied to the PSD circuits,one compensation circuit could not compensate for both detectors. Twocompensation circuits would be required, one for each PSD. This, ofcourse, would increase significantly the circuit complexity of theinduction tool and a reduction in its overall reliability.

It is a characteristic of induction tools that at low conductivities,the amount of direct mutual coupling ("X" sonde error) between thetransmitter coil and the receiver coil, even in a tool which employs asystem of receiver coils which minimize this mutual coupling, is notzero. In fact, a ratio of 10:1 of the signal response due to directmutual coupling to the R component in the receiver signal is notuncommon. When encountering low conductivities, in order to resolve theR component to ±1% accuracy, a phase accuracy of 1 milliradian isrequired. For the case of high conductivities, the R component willexceed X by a factor which can be substantial, i.e., "R"=10×"X". Forthis case, to resolve X to ±1% would likewise require a high degree ofphase accuracy.

To obtain accurate phase component signal measurements that areessentially free of the static and temperature dependent phase shifterrors, a highly phase stable, low distortion transmitter signal must begenerated. A highly phase stable transmitter signal is required toinsure phase accuracy between the signals of the tool in the generationof the transmitter signal and in the detection of the phase componentsignals in the receiver signal. The requirement for low distortion inthe transmitter signal results from the frequency response of theearth's formations.

A known phenomenon in induction logging is the difference in theformation response as a function of frequency and formationconductivity. In general, the response signal received by an inductiontool at low conductivities increases as the square of the frequency fora constant transmitter current. Because of the greater formationresponse at higher frequencies than at lower frequencies over most ofthe conductivities encountered, it becomes apparent that a lowdistortion transmitter signal is required. The more distorted thetransmitter signal is, the larger in amplitudes are the harmonics of thefundamental frequency. Such harmonics propogate through the formationfrom transmitter to receiver with an attenuation and phase shift notrelated to those of the fundamental frequency. They can thus introducefalse signals into the receiver that may cause a misleading result to beobtained from the induction tool measurement. Thus, more noise will bepresent in the resulting receiver signal from these higher frequencyharmonics.

This variation in formation response with frequency can be put to gooduse to extend the range of formation resistivity that may be accuratelymeasured by an induction logging tool. At high formation conductivitiesand higher frequencies, a phenomenon known as "skin-effect" causes aloss of proportionality between the received signal and formationconductivity, introducing additional complexity in the interpretation ofthe signals.

Additionally, at the lower transmitter frequencies and at lowconductivities, the response from the formation falls below the noiselevel of the induction logging system. In this case, meaningfulmeasurements are impossible. Thus, when encountering low conductivities,a high frequency for the transmitter signal would provide the moreaccurate reading of the formation conductivity. But, because of thesloping away of the response curves for the higher frequencies at higherconductivities, it would be desirable to have a lower transmitterfrequency at high conductivities to avoid ambiguity in the conductivityderived from those measurements. This may be achieved by selection of asingle frequency appropriate for the conductivity range expected priorto logging, or by the generation of two or more frequenciessimultaneously in the transmitter, with subsequent frequency separationin each receiver circuit and in each phase selective detection circuit,or by sequentially switching to different frequencies while logging.

Yet another problem present in prior-art logging tools has been theproblem of determining from the measured tool output responses the trueand correct characteristic of the formation. That is, determining thetransfer function of the tool relating the tool input signal,representative of the formation characteristic, to the measured tooloutput response. It is from this transfer function that the true valueof the formation characteristic is inferred based on the measured outputresponses.

Because of variations in circuit parameters as a result of temperaturechanges, (e.g., changes in the amplifier gains) the calibrated transferfunction of the tool at one operating position may not be the same as atanother. A determination of the transfer function is normally effecteduphole by placing one or more signal sources near the receiver coil tosimulate various formation conductivities. The responses to these testsignals are recorded and used to derive a calibration transfer functionfor the tool. This function is thereafter used as the function relatinginput to output of the logging tool. Yet, for prior-art tools, the dataobtained during a logging run is not corrected for the effects oftemperature changes, during logging, to the transfer function.

A further characteristic of all induction logging tools is the very widedynamic range present in the detected phase component signals over whichuseful information is contained. A dynamic range of 10,000:1 (>80 db) isnot uncommon. Superimposed on the useful information in a detectedcomponent signal is a certain amount of random noise which degrades thequality of any measurements made. In analog prior-art induction loggingtools (as distinguished from a digital logging tool), this noiseincludes noise generated during the transmission of the detected analogphase component signals to the surface through a wireline logging cable.Analog transmission of the phase component signals uphole is subjectedto the problem of signal degradation by the introduction of errorpotentials and noise or cross talk in the electrical leads of thelogging cable.

Prior-art logging tools have attempted to handle the large dynamic rangein the detected component signals in different ways. U.S. Pat. No.3,056,917 discloses one such technique in which the dynamic range isdivided into two parts--a first range in which the transmitter currentis adjusted to obtain a constant receiver signal voltage and a secondrange in which the transmitter current is held constant. A signal isthen recorded which is representative of the transmitter current whenthe receiver signal is constant, and which is representative of thereceiver signal when the transmitter current is held constant. Theresulting recorded signal represents the conductivity of the formationin the first range and the resistivity of the formation in the second.Yet other prior-art techniques for handling this large dynamic range inthe detected phase component signals are also discussed in U.S. Pat. No.3,056,917.

Most prior-art tools have used standard techniques to try to eliminateor minimize the amount of noise introduced into the analog signalstransmitted over the logging cable. The use of twisted wire pairs,shielded leads, low noise slip rings, etc. are but a few. Where aninduction tool requires precise, accurate measurements of the detectedsignals, regardless of their magnitudes, these prior-art techniques areno longer adequate.

Because of the limitations present in the prior-art logging tools andthe need for more precise and accurate measurements of the phasequadrature components of the receiver signal, it would be advantageousto provide an induction logging tool to measure and convert to digitalform downhole the wide dynamic range in the detected phase componentsignals, and to measure them with the same resolution and accuracy atall levels of signals. These digital signals are subsequentlytransmitted to the surface substantially uncorrupted by noise aspreviously discussed. It would also be advantageous to dynamicallycompensate for both the static and temperature dependent phase shifterrors in the circuits of the tool involved in the generation of theformation currents and in the detection of the phase components of thereceiver signal.

It would also be advantageous to provide an induction tool whichdigitally generates downhole both a highly phase stable, low distortiontransmitter signal and a highly stable phase reference signal in orderthat a single phase sensitive detector may sequentially detect both theR and the X phase quadrature component signals while compensating forthe phase shift errors. It would also be advantageous to provide adigital induction logging tool in which the frequency of the digitallygenerated transmitter signal is selectable from among a plurality oftransmitter frequencies. It would also be advantageous to provide aninduction tool which automatically selects, during a logging run, thetransmitter frequency or frequencies that will produce the optimumformation response signals for the conductivities actually encounteredby the tool. It would also be advantageous to provide an inductionlogging system which automatically produces, during a logging run, testcalibration measurements which are used to derive a linearizationcorrection function to correct for temperature dependent variations inthe transfer function of the tool at any time during the logging run.

SUMMARY OF THE INVENTION

In accordance with the present invention, a digital induction loggingtool is provided for measuring a characteristic of the earth'ssub-surface formations by causing formation currents to flow in responseto a transmitter signal of a predetermined frequency and by measuring areceiver signal generated in response to these formation currents. Thedigital induction tool obtains digital floating-point samples of phasequadrature components of the receiver signal at various depth pointsalong the borehole. These samples are obtained by successively measuringin one phase sensitive detector circuit the component signal in-phasewith the transmitter signal and the component signal in quadraturethereto. Means are included in the tool for automatically compensatingfor the phase shift errors in the phase quadrature componentmeasurements introduced by circuits of the tool.

A surface located central processing unit (CPU) is programmed to producecharacteristic data of the earth's sub-surface formations from thefloating point digital samples obtained by the digital induction tool.The CPU transmits command information to the digital induction loggingtool downhole to specify the operating modes and parameters forobtaining the floating point digital samples. A digital telemetry meansis used to transmit the digital information between the inductionlogging tool and the CPU over a wireline cable suspending the inductiontool in the well borehole.

The digital induction logging tool includes a transmitter coil thatresponds to a low distortion, phase stable sinusoidal transmitter signalto induce a magnetic field into the earth's sub-surface formations. Thismagnetic field causes eddy currents to flow in the formations. Theseeddy currents themselves produce magnetic fields. A receiver coilresponds to the magnetic fields generated by the formation currents togenerate a receiver signal indicative of a characteristic of theformations, i.e., the formation conductivity. A controller is includedfor controlling the internal timing and functional operations of thetool's circuitry. The controller responds to the digital command andcontrol signals transmitted from the CPU located at the surface.

Also included in the digital induction logging tool is a waveformgenerator responsive to the controller for digitally generating a lowdistortion, phase stable sinusoidal transmitter signal from among atleast two selectable transmitter frequencies. Selection of thetransmitter frequency may be automatically controlled by the CPU, orunder operator control where selection is made to obtain the moreaccurate reading of the characteristic of the earth's formations.Selection may be based upon the current value of the conductivity of theformations being encountered.

The waveform generator includes a read-only-memory which containsdigital information representative of magnitude values of thetransmitter signal to be generated. The read-only-memory responds to anaddress counter to output digital code words to a digital-to-analogconvertor to generate a stair-step approximation to the desiredsinusoidal transmitter waveform. The waveform generator also includes afilter connected to the output of the digital-to-analog converter forsmoothing the stair-step sinusoidal waveform by filtering the harmonicstherefrom. A transmitter amplifier amplifies the filtered sinusoidalwaveform to obtain the low distortion sinusoidal transmitter signal thatis actually applied to the transmitter coil. In one embodiment of theinvention, the waveform generator is programmed to produce a waveformconsisting of two different superimposed sinusoidal frequencies allowingfor simultaneous multi-frequency logging.

The digital induction logging tool also includes an autophase unit thatresponds to the controller to generate a digital phase reference signalto a phase sensitive detector for resolving the receiver signal into itsinphase and quadrature components. The phase reference signalsuccessively changes from a first to a second phase relationship withthe transmitter signal in response to commands from the controller. Thefirst and second phase reference signals respectively having the firstand second phase relationships being precisely orthogonal to oneanother. The autophase unit further includes first and second flip-flopsinterconnected such that the output signal from the first flip-flop (thephase reference signal) is phase shifted from the first phaserelationship to the second phase relationship in response to controlsignals from the controller.

The digital induction logging tool also includes a phase sensing meansresponsive to the receiver signal and the phase reference signal fromthe autophase unit for successively detecting the phase quadraturecomponents in the receiver signal. Each detected component is thatcomponent in-phase with the current phase of the phase reference signal.Included in the phase sensing means is the phase sensitive detector anda receiver amplifier for amplifying the signal from either the receivercoil or a test signal derived from the current in the transmitter coil.The phase sensing means also applies a feedback error signal to theautophase unit. The feedback error signal represents the magnitude ofthe detected component signal generated during an autophase cycle.

During each autophase cycle, a phase reference signal is generated fordetecting the quadrature component of the receiver signal. Alsogenerated during each autophase cycle is a test signal derived from thetransmitter current. Based on the phase relationship between the testsignal and the phase reference signal generated during an autophasecycle, the feedback error signal causes the autophase unit to phaseshift the phase reference signal in a direction to reduce to zero thedetected reactive component. The amount of phase shift applied by theautophase unit is retained at the completion of each autophase cyclethereby to compensate for phase shift errors introduced by circuits ofthe tool.

Also contained in the read-only-memory of the waveform generator arereference clock generating signals that are output along with theamplitude data. The reference clock signal is applied to the autophaseunit and is used to generate the phase reference signal. The referenceclock generating signals are stored in the read-only-memory relative tothe transmitter signal generating data so that the resulting phasereference signal from the autophase unit is phase shifted relative tothe transmitter signal to compensate for phase shift errors introducedby circuits of the tool.

The digital induction tool also includes a floating pointanalog-to-digital converter for successively obtaining floating pointdigital samples of the magnitude of the detected phase quadraturecomponents of the receiver signal output from the phase sensing means.Each floating point signal includes a digital word signal representingthe exponent of a floating point number and a digital word signalrepresenting its magnitude. The floating point analog-to-digitalconverter includes a voltage-to-frequency converter that generates adigital clocking frequency proportional to the magnitude of thecomponent signal output from the phase sensing means. A counter countsclock cycles of the digital frequency signal during a predetermined timeperiod. This predetermined time period represent an integration timeover which the detected component signal is be integrated.

A shift register having a stage for each bit of the counter receives andstores the contents of the counter at the end of each predetermined timeperiod. Responsive to shift pulses, the shift register shifts theresulting count in a direction to increase the magnitude of the countcontained in a predetermined sub-set number of output bits of the shiftregister. This sub-set of bits has a sign bit and most significant bit(MSB), and forms the magnitude of each floating point digital signal. Asecond counter is also included for counting the number of shift pulsesapplied to the shift register. The shift register is shifted one bitposition for each shift pulse. This shifting continues until the firstto occur of either a logic 1 signal appearing in the MSB of themagnitude or the count in the second counter reaches a predeterminedcount (zero). The count in the second counter forms the exponent of eachfloating point digital sample.

BRIEF DESCRIPTION OF THE DRAWINGS

A digital induction logging tool constructed in accordance with thepreferred embodiment of the invention is illustrated in the accompanyingdrawings in which,

FIG. 1 is an illustration of a digital induction logging system in placefor logging a borehole;

FIG. 2 is a functional circuit block diagram of the digital inductionlogging tool illustrated in FIG. 1;

FIG. 3 is a phase diagram illustrating how the voltage induced in thereceiver coil system varies as a function of the transmitter currentfrequency;

FIG. 4 is a phase diagram illustrating the phase relationships betweenthe current in the transmitter coil system, the transmitter voltageoutput, and the R and X signals received in the receiver coil system;

FIGS. 5A and 5B are a more detailed circuit diagram of the controllerillustrated in FIG. 1;

FIG. 6 is a timing diagram for various signals of the floating point A/Dconverter illustrated in FIGS. 2 and 12;

FIG. 7 is a flow diagram of the controller firmware routine forgenerating a time interval;

FIGS. 8A and 8B are a firmware flow diagram for the firmware of thecontroller shown in FIGS. 5A and 5B;

FIGS. 9A and 9B are a more detailed circuit diagram of the digitalwaveform generator which generates the transmitter signal;

FIG. 10 is an illustration of the phase relationship between thetransmitter signal voltage waveform and the phase reference signalapplied to the phase sensitive detector for detecting both the R and theX phase component signals in the receiver signal;

FIG. 11 is a more detailed circuit diagram of the autophase unitincluded in FIG. 2;

FIG. 12 is a more detailed circuit diagram of the floating pointanalog-to-digital converter shown in FIG. 2;

FIG. 13 is an illustration of the autocalibration linearizationtechnique; and

FIG. 14 is a graph of the receiver signal versus formation conductivityfor different transmitter frequencies.

Similar reference numerals refer to similar parts throughout the severaldrawings.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT OF THE INVENTION TheDigital Induction Logging System

Referring now to the figures and first to FIG. 1, a pictorialrepresentation of a digital induction logging system including thepresent invention is shown. A digital induction resistivity logging tool1 in accordance with the invention is shown suspended in a well borehole16 by a wireline cable 14. Associated with each end of the wirelinecable 14 are telemetry transmitter-receiver units 12 which togethercomprise a telemetry means for transmitting and receiving digitalinformation between a surface located central processing unit (CPU) 10and the downhole digital induction tool 1. Telemetrytransmitter-receiver units 12 operate to convey command and datainformation from the CPU 10 to the circuits of the digital inductiontool 1, and to transmit and receive the floating point digital signalsobtained by the digital induction tool 1. These digital samplesrepresent a characteristic of the earth's sub-surface formations and aretransmitted uphole to the CPU 10 for further processing.

By means of a suitable drum and winch mechanism (not shown) the lengthof cable which is suspended in the borehole may be either increased ordecreased to provide the desired movement of the downhole apparatusthrough the borehole.

The Digital Induction Logging Tool 1

Turning now to FIG. 2, a more detailed block diagram of the digitalinduction logging tool 1 is shown. The induction logging tool operateson principles that are well known to those skilled in the art and willonly be briefly described herein. A transmitter coil 34 is excited witha AC varying transmitter current i_(T). The presence of this transmittercurrent i_(T) produces a magnetic field which propagates into theearth's sub-surface formations surrounding the induction tool 1 sonde.This magnetic field induces eddy currents to flow in the formations.Positioned proximate the transmitter coil 34, but electrically isolatedfrom direct coupling of the magnetic field present in the transmittercoil 34, is a receiver coil 36. As a result of eddy currents flowing inthe formations, magnetic fields are, in turn, generated. These magneticfields are detected by the receiver coil 36. A receiver signal voltage37 (V_(r)) is thus generated on the output of the receiver coil 36. Thisvoltage is indicative of the conductivity of the formations. From thisreceiver signal 37, phase components will be obtained for furtherprocessing by the CPU 10 to obtain the desired characteristic of thesub-surface formations.

It should be pointed out that a single transmitter coil and a singlereceiver coil are shown in FIG. 2 for purposes of illustration anddiscussion of the invention. Although the invention is described withreference to single transmitter and receiver coil systems, the inventionis equally applicable to systems which include multiple transmittercoils or multiple receiver coils or combinations thereof. For an exampleof an induction logging systems which incorporates multiple coilarrangements, see U.S. Pat. No. 3,150,314. Particular benefits areobtained from such coil systems. For example, modern coil arrays aredesigned to substantially eliminate receiver voltage signal due tomutual inductive coupling (X sonde error).

The circuits illustrated in FIG. 2 perform their functions under controlof a controller 22. The telemetry transmitter-receiver unit 12 respondsto the telemetry bus data from the wireline cable 14 to apply thecommand and data information from CPU 10 to controller 22. This dataspecifies the modes and parameters of the induction tool. A quartzcrystal controlled clock oscillator 24 provides the master system timingsignal 25 that is applied to the controller 22 via a digital waveformgenerator 26. Digital waveform generator 26 divides clock signal 25before applying it to the controller 22. From clock 24 is derived all ofthe circuit timing signals. Synchronization and control of the variousfunctional blocks depicted in FIG. 2 by controller 22 is described inmore detail below.

Still referring to FIG. 2, a waveform generator 26 responds to thesystem clock 25 to produce a stair-step approximation to a sinusoidalwaveform on lead 27 which ultimately will become the AC transmittersignal on lead 28 applied to the transmitter coil 34. The frequency ofthe transmitter signal is selectable from among a plurality offrequencies which waveform generator 26 is capable of generating.Frequency select signals may be supplied from the surface at any timeduring downhole operations to select the desired transmitter frequency.In addition to generating a stair-step approximation to the transmittersignal, waveform generator 26 also provides several clocking signals toan autophase unit 32. Autophase unit 32 has two primary functions:First, to generate the phase reference signal 42 to a phase sensitivedetector 40. The phase reference signal 42 enables that detector todetect the phase quadrature component signals of the receiver signal 37.Second, to phase shift the phase reference signal 42 in a direction tominimize the output signal from the phase sensitive detector 40 during aphase compensation cycle. Minimizing the output of detector 40eliminates phase shift errors introduced by certain circuits of thetool.

Connected to the output of the waveform generator 26 is a low passfilter 86 which filters the harmonic content of the stair-stepapproximation signal on lead 27. The output of low pass filter 86 isthen applied to a transmitter power amplifier 88 which amplifies andapplies a sinusoidal transmitter signal on lead 28 to the transmittercoil 34. Connected between the output of transmitter power amplifier 88and circuit ground is a capacitor C1 which functions to apply a powerfactor adjustment to the transmitter signal on lead 28. Two additionalcapacitors, C2 and C3 may be connected in parallel with capacitor C1 viasolid state switches responding to the frequency select signals F₀ andF₁. Since one of the features of the digital logging systems accordingto the present invention is the capability of selecting a transmitterfrequency from among a plurality of transmitter frequencies (describedin more detail below), capacitors C2 and C3 are provisonally provided toprovide the additional power factor compensation to the output of thetransmitter power amplifier 88 as a function of the selected transmitterfrequency, and thus reduce the power consumption (and heat dissipation)of the transmitter.

Connected in series with the output of the transmitter power amplifier88 is a primary winding of a current transformer 31. Also connected inseries with the primary winding of transformer 31 is the transmittercoil 34. The transmitter current i_(T) also flows in the primary windingof current transformer 31. Connected across the secondary winding oftransformer 31 is a parallel combination of resistor R and capacitor C8.The voltage signal developed across R functions as a test referencevoltage sampled from the current flowing in the transmitter coil 34.From this reference signal will come the test signals used in autophasecompensation for phase shift errors introduced by measurement circuitsin the tool and for autocalibrating the transfer function of the toolCapacitor C3 function to apply a small amount of phase shift to thereference test signals generated across R. This phase shift is intendedto duplicate the phase shift that is present in V_(r) at the output ofthe receiver coil 36 due to the imperfectness of the coils 34 and 36.

Connected across R and C is the primary winding of transformer 29. Thesecondary of transformer 29 has a plurality of output taps labeled a, b,c and d, each tap generating a different voltage level. A controllableswitch S2 selects from among the output tap point of transformer 29 toobtain the test signal 33.

Still referring to FIG. 2, receive coil 36 is shown with the thereceiver voltage V_(r) at its output terminals. As previously discussed,V_(r) is generated in response to the magnetic fields produced by theeddy currents flowing in the formations. In normal operation, thereceiver voltage on lead 37 is applied through a controllable switch S1as the input to a receiver amplifier 38. The output of the receiveramplifier 38 is applied as the input voltage to the phase sensitivedetector 40. Phase sensitive detector 40 and preamplifier 38 comprise aphase sensing means 43 that is used to detect the quadrature phasecomponent signals, R and X, in the receiver signal V_(r).

In order for phase sensitive detector 40 to detect these phase componentsignals, a phase reference signal 42 having the same phase as the phaseof the component to be detected must be generated and applied to thephase sensitive detector 40. The phase relationship between the phasereference signal 42 and the current in the transmitter coil 34 (i_(T))determines which phase component of the receiver signal will bedetected. For the present invention, a single phase sensitive detector40 sequentially detects both quadrature phase components of the receiversignal, i.e., the phase component that is in-phase with i_(T) and thecomponent that is orthogonal thereto. For phase sensitive detector 40 tosequentially detect both quadrature components, it is necessary tosequentially generate the phase reference signal 42 alternately havingtwo phase relationships to i_(T). First, a phase relationship that willproduce the in-phase phase component signal R, and second, a phaserelationship that will produce the orthogonal phase component X. Toinsure orthogonality between the R and X components, the phase changebetween the first and second phase relationships must be precisely 90°.

The autophase unit 32 responds to clocking signals from waveformgenerator 26 and to the controller 22 to produce the phase referencesignal 42 having the sequential phase relationships to the transmittercurrent i_(T). Waveform generator 26 outputs, in a predetermined phaserelationship to the generated sinusoidal transmitter waveform on lead27, a referenced clock signal on lead 50 (4f) that is used by autophaseunit 32 to generate the square wave phase reference signal 42. A moredetailed description of the circuitry of the autophase unit 32 is givenbelow.

Still referring to FIG. 2, two outputs are produced by the phasesensitive detector 40, an analog signal on lead 41 representing thedetected phase component signal in the receiver signal 37 and a feedbackerror sighal AUTOφ on lead 45. The detected phase component signal isapplied to a floating point analog-to-digital convertor 46 while AUTOφis applied to the autophase unit 32. The signal AUTOφ is a feedbackdigital signal indicative of the polarity of the detected phasecomponent signal. This feedback error signal functions as part of aclosed loop control system that is used to adjust the phase of the phasereference signal 42 to compensate for variable phase shift errors(mostly temperature dependent) of the circuits of the tool involved inthe generation of the transmitter signal and in the detection of thephase component signals in the receiver signal on lead 37.

Ideally, if the circuits of the tool transmitter and receiver coils anddownhole electronics were ideal (not subject to temperature drift, allinductors were pure inductance, etc.), the phase of the components inthe receive signal 37 would be predictable and constant. Unfortunately,such an ideal world does not exist. As a result, phase shift errors areintroduced into the various signals of the tool that are notpredictable. Basically, two different sources of phase shift errors arecompensated for with the apparatus according to the invention. Thosephase shift errors introduced by the circuits of the downhole coils andelectronics when measured at a fixed temperature and operating conditionare defined as "the static sonde phase shift errors." Those phase shifterrors introduced by such things as temperature drift are defined as"dynamic phase shift errors." These dynamic phase shift errors act tomodulate the static phase shift errors. In the present invention, thesestatic phase shift errors are compensated for by a predetermined phaseshifting of the referenced clock signal on lead 50 relative to thedigitally generated transmitter signal on lead 27, both signalsgenerated by the waveform generator 26. As is more fully discussedbelow, the digital data used to generate both the referenced clocksignal on lead 50 and the transmitter signal on lead 27 are stored in aread-only-memory relative to one another such that the desired phaseshift to compensate for these static phase shift errors result when thecontents of the memory locations are read out. In a sense, the phaseshifting of the referenced clock signal on lead 50 relative to thetransmitter signal on lead 27 comprises a first order phase shiftcorrection to the total phase shift error that is present in the coiland electronic circuits, both static and dynamic. The static phase shiftcorrection is also adapted to the frequency selected for thetransmitter, because the coils and electronics introduce a frequencydependent phase shift.

The autophase unit 32 includes a means for phase shifting the phasereference signal on lead 42 by phase shifting the reference clock signalon lead 50 from which the phase reference signal on lead 42 is derived.Normal operation of the digital induction tool according to theinvention is interrupted periodically by an autophase cycle during whichcorrections to the phase relationship of the phase reference signal onlead 42 and the transmitter current i_(T) are made to compensate for thetemperature dependent phase shift errors present in the phase sensingmeans 43. This is in effect a second order phase shift errorcompensation.

During each autophase cycle, controllable switch S1 which is part ofselecting mean 47 is controlled by the controller 22 to select the testsignal on lead 33 which is derived from the transformer 31. This testsignal on lead 33 is applied as the input signal to the receiveramplifier 38, and functions as a reference signal with a known phaserelationship to the transmitter current i_(T). Responsive to the commandSTART AUTOPHASE from the controller 22, autophase unit 32 generates aphase reference signal 42 having an approximately 90° phase relationshipto the test signal 33. If there was no change in the phase shiftintroduced by the phase sensing means 43 from the last autophase cycle,the detected output phase signal on lead 41 should be at or near zero.For such a condition, no further phase shifting to the phase referencesignal 42 is needed in this autophase cycle. However, if the signal onlead 41 is not zero, AUTOφ will generate a feedback error signal to theautophase unit 32 enabling the phase shifting means 95 contained therein(refer to the discussion with reference to FIG. 11) to adjust the phaseof the phase reference signal 42 in a direction to minimize or reducethe magnitude of the detected phase signal on lead 41 back to zero. Atthe end of each autophase cycle, autophase unit 32 retains the amount ofphase shift applied to the phase reference signal on lead 42 until thenext autophase cycle. Thus, temperature dependent drifts, and the like,in phase shift error introduced by the phase sensing means 43 will bedynamically compensated for.

As shown in FIG. 2, the detected phase component signal produced by thephase sensitive detector 40 is applied to a floating pointanalog-to-digital converter 46. The resulting digital samples are thentransmitted via the telemetry transmitter-receiver unit 12 to thesurface located CPU 10. A detailed description of the circuits andoperation of the converter 46 is given below with reference to FIG. 12.

The Controller 22

Turning now to FIG. 5, a detailed circuit diagram of controller 22 isshown. Controller 22 is a general purpose stored program controller inwhich firmware routines are contained in a read-only memory ROM 57.Controlling the internal timing sequencies of the controller 22 is astate controller 50 that responds to a 2.5 MHz clock signal fromwaveform generator 26 to produce the various state timing signals shownin FIG. 6. These timing signals are used to control the sequencing ofthe circuits illustrated in FIG. 5. Among other functions, controller 22operates to generate control pulses on a number of output lines; setslogic levels in a number of input lines and responds to these signals;generates variable time delays using an internal counter which respondsto a clock developed from the system clock; transfers 8-bit wordsdirectly from memory to external circuits; jumps to arbitrary programlocations within the firmware ROM 57 under program control or using anexternally generated 8-bit address (vectored jumps); and, performssimple boolean operations on two binary bits. With these functions, thecontroller 22 controls the internal operations of the induction toolthrough an orderly sequence of events, such as timed measurements;complex serial bit operations and floating-point conversions; and, thegeneration of control pulses for telemetry interface tasks in thetelemetry transmitter-receiver unit 12.

A counter 56 consisting of an 8-bit up counter (two 4-bit up countersconnected in series) with parallel load input is used to implement theprogram counter function. Program counter 56 addresses ROM 57 to accesseach program instruction. When the controller encounters a "jump"instruction, the new program counter value is loaded into counter 56 viathe D0-D3 data input lines of the two 4-bit counters.

The heart of the controller 22 is an instruction decoder unit 52 which,for the presently preferred embodiment of the invention, is a MotorolaMC-14500 Industrial Control Unit (ICU) which is described in detail inthe publication "Motorola MC-14500B Industrial Control Unit Handbook,"authored by Vern Gregory and Brian Dellande and published in 1977 byMotorola, Inc. The ICU 52 recognizes a set of only 16 instructions.

A program is stored in ROM 57 in the form of alternate instructions andaddress words which are output on the controller data bus 65. Thisinterleaved structure allows for higher operating speed and a reductionin the number of interconnection wires. The first 4-bit nibble of each8-bit instruction word is accepted by the ICU 52, and the second nibbleforms an address to a one-of-16 decoder unit 64 to generate a pulse onone of 16 output lines. The succeeding address word in each programinstruction is used to define an input or output port through which thecontroller communicates with an external circuit. Table 1 belowillustrates the program instruction storage for the ROM 57.

                  TABLE 1                                                         ______________________________________                                                Output Bits from ROM                                                                              Program                                           ROM Address                                                                             D7    D6    D5  D4  D3   D2  D1  D0   Location                      ______________________________________                                        0         ICU instruction                                                                           Decoder 64 address                                                                          1                                         1         data                                                                2         ICU instruction                                                                           Decoder 64 address                                                                          2                                         3         data                                                                4         ICU instruction                                                                           Decoder 64 address                                                                          3                                         5         data                                                                6                               4                                             ______________________________________                                    

A "read" or "write" instruction allows the ICU 52 to obtain one binarybit through the input multiplexer 60 or to load one binary bit in anoutput register via addressable latches 62 and decoder 64. Programmabletime delays are generated by controller 22 by loading a 16-bit downcounter 58 with two 8-bit data words. Counter 58 consists of two 8-bitdown counters connected in series, counter A and counter B. The periodcan be set from 2⁰ to 2¹⁶ major clock cycles, corresponding to a rangeof 3.2 microseconds to 209 milliseconds with an input clock of 2.5 MHz.One instruction that is particularly useful in the controller 22 allowsfor an external circuit (such as a counter) to control the data bus andthereby to load a new number into the program counter, giving a"vectored jump." This is equivalent to an interrupt in the normaloperation of the stored program.

Referring now to FIG. 7 which illustrates the program flow diagram togenerate a specified time delay T by controller 22, the count number, C,representing the number of clock cycles to be counted by down counter 58must be separated into two bytes destined for each half of the counter58. Counter A handles the most significant byte and counter B (whichmust be loaded first) the least significant. The output from thecounters is a zero detect bit ZD which is fed back to the X1 input ofmultiplexer 60. The zero detect bit is obtained from counter A (mostsignificant byte). To obtain the correct count, this bit must go low(indicating that counter A is empty) and then go high (indicating thatboth counters A and B are empty). Both states must be detected to definethe instant of the rising transition or the point at which both countersare at zero. The following is an example of the programming of ROM 57 togenerate a time delay of 80 milliseconds with a 2.5 MHz system clockaccording to the flow diagram of FIG. 7: First, the counter 58 must beloaded with the number of clock cycles to be counted.

    ______________________________________                                        Program  Machine                                                              Location Code           Action                                                ______________________________________                                        20       01             load counter B                                                 A8             least significant byte                                21       02             load counter A                                                 61             most significant byte                                 ______________________________________                                    

The zero detect bit ZD must now be loaded and tested. This uses the ICUinstructions "load I/0 bus into result register (RR)" and "skip if RR iszero", if not "loop back." The ZD information is available on input port1 with an address of 01 and the instruction code "1".

    ______________________________________                                        Program   Machine                                                             Location  Code           Action                                               ______________________________________                                        22        IF             I/O bus to RR                                                  01             input port 1 (ZD)                                    23        EF             skip if RR = O                                                 00                                                                  24        CF             jump to program                                                22             location 22                                          ______________________________________                                    

The machine will sit in this loop until the counters reach zero, atwhich time the instruction at location 24 is skipped and the routine iscontinued. Next, the zero detect bit is tested until it goes high usingthe same operation as described above except that the complement of thezero detect bit is loaded:

    ______________________________________                                        Program  Machine                                                              Location Code         Action                                                  ______________________________________                                        25       2F           complement of I/O bus                                            01           input to RR 1                                           26       EF           skip if RR = O                                                   00                                                                   27       CF           jump to program                                                  24           location 25                                             ______________________________________                                    

When the ZD bit of counter A again return to a logic high, both counterswill then be at zero and the desired time interval will have beengenerated and the program will jump out of the loop to the nextinstruction following location 27.

Illustrated in FIG. 8 is the firmware program flow diagram for thepreferred embodiment of the present invention. Shown at various pointsof the diagram are "time out" blocks which represent a variable timedelay generated in the same manner as the above described example.

The Waveform Generator 26

As previously discussed with reference to FIG. 2, the digital waveformgenerator 26 produces a stair-step approximation to a sinusoidalwaveform. This waveform is then filtered by a low pass filter 86 andamplified in a power amplifier 88 to produce a low distortion, highlystable sinusoidal transmitter signal that is applied to the transmittercoil 34. The frequency of the transmitter signal is selectable fromamong a plurality of transmitter frequencies, e.g., 10 KHz, 20 KHz, 40KHz, which waveform generator 26 is capable of generating.

A low distortion transmitter signal is desirable because, as will beseen below, the amplitude response of the receiver coil increases as thesquare of the transmitter frequency. Accordingly, the higher theharmonic content, the greater the distortion will be in the receivedsignal.

Illustrated in FIG. 3 is a vector diagram showing the phase relationshipbetween the transmitter current i_(T) and the voltage that is inducedinto the formation and the voltage that is induced into the receivercoil 36 for various transmitter frequencies. It has been appreciated forsome time by those skilled in the art that improved measurements in theresistivity of a high conductivity formation (where skin-effectphenomenon significantly effects the response) would be achieved atlower transmitter frequencies (on the order of 10 KHz) while an improvedmeasurement of low conductivity formations (where skin-effect does notsignificantly effect the response) can be improved by raising thefrequency (on the order of 40 KHz) because the amplitude response of thereceiver signal, or sensitivity, of an induction tool varies as thesquare of the frequency of the transmitter current. As used herein,"sensitivity" is defined to be the receiver coil 36 voltage divided bythe transmitter coil current at a given formation conductivity andfrequency, e.g., 10 uv/A at 1 mmho and 20 KHz. Because of this increasein receiver signal with increased frequency, the need for low harmoniccontent in the transmitter signal is easily recognizable in order tominimize distortion in the receiver signal V_(r). A further need for lowdistortion results from the fact that the harmonics of the fundamentalfrequency transmitted propogate through the formation with anattenuation and phase shift not related to those of the fundamentalfrequency. They can thus introduce false signals into the receiver thatmay cause a misleading result to be obtained from the induction toolmeasurement.

In addition to digitally generating the low distortion, phase stabletransmitter signal on lead 28, the digital waveform generator 26 alsoproduces a reference clock signal on lead 50 that is applied to theautophase unit 32. The clock on lead 50 is used by autophase unit 32 togenerate the phase reference signal on lead 42 to phase sensitivedetector 40 to detect the desired phase components of the receive signalon lead 37. Additionally, the phase reference clock on lead 50 isgenerated with a phase shift relative to the generated transmit signalon lead 28 (see FIG. 10) so that the static sonde phase shift errors, aspreviously defined, are compensated for. In other words, by phaseshifting the clock on lead 50, the detected phase component signal onlead 41 will not contain static phase shift errors introduced by thecircuits involved in the generation of the transmitter signal and in thedetection of the phase component signals in the receiver signal V_(r) onlead 37.

The description of the method by which the digital waveform generator 26produces the low distortion, highly phase stable transmitter signal andthe phase shifted reference clock signal on lead 50 can best beunderstood by referring now to FIG. 9. FIG. 9 shows a detailed circuitdiagram of the digital waveform generator 26. Also illustrated is thecircuit for the crystal controlled clock 24 shown in FIG. 2. The outputof clock 24 is applied to a 12-bit binary counter 66. Counter 66functions both as the ROM 70,72 address generator, and as a clock signalgenerator to the controller 22 (2.5 MHz) and to autophase unit 32 (SARCLOCK).

Generation of the digital stair-step approximation waveform on lead 27and the generation of the reference clock signal on lead 50 arespecified by digital code words stored in ROM memory chips 70,72. Thesetwo memory chips form a 512×16-bit ROM. The output from the 12-bitbinary counter 66 is used to address ROM 70,72 to output these storeddigital code words. A set of exclusive OR gates 76 respond to the outputfrom ROM 70,72 to generate digital codes that represent magnitude valuesof the transmitter signal to be generated. These digital code words areapplied to a digital-to-analog convertor 79 containing input latches78,80 that store, for one clock cycle, the magnitude code words. An R-2Rprecision resistor network 82 responds to the output from latches 78,80to generate an analog voltage according to each stored digital codeword. Operational amplifier 84 responds to the output current of theprecision resistor network 82 to generate an output analog voltage. Asthe address counter 66 cycles through its addresses, the stair-stepwaveform on lead 27 (see also FIG. 10) is produced on the output of thedigital-to-analog convertor 79. This signal is filtered in low passfilter 86 and amplified in power amplifier 88. The output from poweramplifier 88 comprises the analog sinusoidal transmitter signal on lead28 that is applied to the transmitter coil 34 to produce the transmittercurrent i_(T).

In addition to the magnitude code words output from ROM 70,72, the ROMalso outputs data to generate the reference clock signal on lead 50. Byappropriately selecting the memory address locations into which thereference clock signal generating data is stored relative to the storedmagnitude code words for the digital stairstep approximation waveform,it is possible to produce a reference clock signal on lead 50 that isphase shifted a predetermined amount relative to the resultingsinusoidal transmit signal on lead 28. By knowing the amount of staticphase shift error that is to be compensated for, the ROM 70,72 can beappropriately programmed.

As previously mentioned, digital waveform generator 26 illustrated inFIG. 9 further includes the capability of generating a transmittersignal 28 having a frequency selected from among a plurality ofpredetermined frequencies. Hex latch 68 is shown in FIG. 9 responding tothe command bus inputs from the telemetry transmitter-receiver unit 12.Commands transmitted from the CPU 10 located on the surface causecertain bits on the output of latch 68 to control two bits of addressingfor the ROM 70,72. These two bits of address control will determine thefrequency of the transmitter signal on lead 28. When the conductivityrange of the formations in a particular location is approximately known,the operator can manually select from among the plurality of possibletransmitter frequencies the transmitter frequency that will give thebest accuracy.

In this regard, FIG. 14 illustrates the relationship between receiversignal and formation conductivity for various transmitter frequencies.As previously discussed, at the lower transmitter frequencies and at lowconductivities, the response from the formation falls below the noiselevel of the induction logging system, making meaningful measurementsimpossible. Thus, when encountering low conductivities, a high frequencyis most desirable. As shown in FIG. 14, at high conductivities, a lowerfrequency is most desirable because of the sloping away of the responsesignal at the higher transmitter frequencies.

In addition to the manual selection of the transmitter frequency, theCPU 10 can automatically select the transmitter frequency which willgive the best measurement based on such parameters as the actualconductivity being measured. For example, the following Table 2illustrates a conductivity-frequency schedule which is advantageous forselecting frequencies according to the invention:

                  TABLE 2                                                         ______________________________________                                        Measured                                                                      Conductivity (mmho/m)                                                                          Frequency (KHz)                                              ______________________________________                                          2 to 10000     10                                                            1 to 5000       20                                                           .2 to 1000       40                                                           ______________________________________                                    

Thus, for high conductivity a low transmitter frequency (10 KHz) couldbe selected to take advantage of the greater linearity of response tohigh conductivity formations, and vice versa (40 KHz) when encounteringlow conductivity. In a third version, the waveform generator isprogrammed to produce a waveform consisting of two differentsuperimposed sinusoidal frequencies. This allows for simultaneousmulti-frequency logging. For such a case, in order to detect theformation response to each transmitter frequency according to thepresent invention, the phase shifting feature of waveform generator 26,autophase unit 32 and the phase sensing means 43 must be duplicated, onecombination for each frequency.

The present invention includes a means in the waveform generator 26 togenerate any one of a plurality of transmitter frequencies or acombination of two or more frequencies in response to digital commandsfrom the CPU 10. As shown in FIG. 3, the three frequencies illustratedfor the transmitter signal for the presently preferred embodiment of theinvention are 10 KHz, 20 KHz and 40 KHz. In order to produce thesefrequencies, with each having the same amplitude and with each generatedfrom the same sequence and rate of addresses from address counter 66,ROM 70,72 has been specially programmed for each frequency. For aspecific sequence of memory addresses, ROM 70,72 generates one-half of acycle of the 10 KHz frequency, one complete cycle of the 20 KHzfrequency and two complete cycles of the 40 KHz frequency. This specificsequence includes the sequence through the possible memory addressesformed by the seven lower order bits, A0-A7, of the 9-bit address forthe ROM 70,72.

The two remaining higher order bits of the 9-bit address, A7-A8, areproduced from the output of latch 68. These two bits represent thefrequency select bits, F₀ and F₁. These two bits specify one of threeaddress sections of the address space for the ROM 70,72. As previouslydiscussed, contained in this address space are both digital magnitudecode words for generating the three possible frequencies and referenceclock generating data. If a 10 KHz frequency is to be produced, both F₀and F₁ will be in a logic 0 state, thus selecting the lower ROM memorylocations. Stored in this ROM memory address space is the generatingdata to produce the one-half of a cycle of the 10 KHz frequency. Thenext higher order address bit from the address counter 66, Q₈, is usedto control one input of exclusive OR gates 76 to cause the digitalmagnitude code words produced on the output of the ROM to becomplemented. In this manner, the second half cycle or negative halfcycle for the 10 KHz frequency can be produced from the same magnitudevalues that were output from the ROM 70,72 to produce the first halfcycle. However, this technique is not applied when the 20 KHz or the 40KHz frequencies are selected.

In the address spaces of the ROM 70,72 for these two higher frequencies,specified by the logic states of F₀ and F₁, the magnitude values storedtherein represent complete cycles of the sinusoidal waveform to beproduced. However, for all three frequencies, the reference clock signalgenerating data is stored relative to the magnitude values in such a wayas to produce the desired phase shift in the clock signal on lead 50 tocompensate for the particular static phase shift errors in the coils andelectronics of the tool at the respective frequency.

The Phase Shift Error Compensation Circuits

Referring again to FIG. 2, the circuits involved in the generation ofthe transmitter current i_(T) and in the detection of the phasecomponent signals in the receiver signal V_(r) can introduce phase shifterrors in the detected phase component voltages. Because the transmittercoil is not a perfect inductor, the phase angle between the transmittercurrent and the induced voltage in the formations differs from the ideal90° phase relationship. This difference amounts to a phase shift errorthat is reflected in the measurements of the phase component signals ifleft uncorrected. Additionally, phase shift errors are introduced by thereceiver coil 36 and the phase sensing means 43 involved in thedetection of the phase component signals themselves. At some steadystate temperature, the phase shift error between the current i_(T) andeither the "R" or "X" phase component signals of the receiver signalV_(r) on lead 37 will be approximately constant. This constant phaseshift error has been defined above as the static phase shift error.Modulating the amount of this phase shift error will be phase shiftsintroduced by such things as temperature variations in the passive andactive component values of the electronics of the phase sensing means43. These phase shift errors are dynamic in nature since the inductionlogging tool according to the present invention is operated in aborehole environment in which the temperature will vary with depth.These phase shift errors have been defined above as the dynamic ortemperature dependent phase shift errors.

The logging tool according to the invention operates to automaticallycompensate for both the static phase shift errors and for thedynamically varying temperature dependent phase shift errors. Staticphase shift errors are compensated for by phase shifting the referenceclock signal on lead 50 relative to the sinusoidal transmitter signal onlead 28. Automatic compensation for the dynamic temperature dependentphase shift errors is achieved during an autophase cycle in which a testsignal is derived from the transmitter coil current i_(T) and is appliedas the normal receiver signal to the phase sensing means 43. During eachautophase cycle, the phase of the phase reference signal on lead 42 isselected to cause the phase sensing means 43 to detect the orthogonalphase component signal in the receiver signal 39 (the "X" component),which, in this case, is the test signal on lead 33. In other words, thephase reference signal on lead 42 during an autophase cycle isapproximately 90° phase shifted relative to the test signal 33 derivedfrom transformer 31.

The feedback error voltage AUTOφ on lead 45, indicative of the polarityof the detected phase component signal during the autophase cycle, isfed back to the autophase unit 32. The signal AUTOφ controls a circuitmeans contained within the autophase unit 32 to phase shift the phasereference signal on lead 42 in a direction to reduce or cause themagnitude of the detected phase component signal during each autophasecycle to approach zero. At the completion of each autophase cycle, theamount of phase shift applied to the phase reference signal on lead 42is retained until the next autophase cycle. In this manner, thetemperature dependent phase shift errors can be dynamically compensatedfor by periodically adjusting the phase of the phase reference signal onlead 42 relative to the transmitter coil 34 current i_(T). Since theautophase unit 32 generates a single phase reference signal on lead 42that switches between two phase relationships to the transmit currenti_(T), and because the phase relationship between the two phase statesis always a precise 90°, it is only necessary to compensate for phaseshift errors at either phase state. This is true because the precise 90°phase shift in the phase reference signal is always obtained regardlessof how much the phase reference signal on lead 42 is phase shifted bythe autophase unit 32.

FIG. 4 illustrates the phase angle relationship between vectors of thecurrent in the transmitter coil 34, the transmitter voltage applied tothe coil(s), and the phase quadrature component signals in the receiversignal V_(r) on lead 37. The phase relationship illustrated in FIG. 4represents a nominal operating condition for the induction tool of thepresent invention in an actual borehole environment. That is, thetemperature for the tool is at the mid-point of the expected temperaturerange that the induction tool is expected to encounter. As previouslydiscussed, the static phase shift errors are compensated for by phaseshifting the reference clock signal on lead 50 relative to the currentin the transmitter coil 34. This phase angle is illustrated in FIG. 4 asθ₃.

Also illustrated in FIG. 4 is the correction range of the autophase unit32 over which autophase unit 32 is able to phase shift the phasereference signal on lead 42 during each autophase cycle. The autophaseunit 32 is able to phase shift the phase reference signal on lead 42,regardless of its phase state, within this correction range tocompensate for the dynamic phase shift errors. Thus, the "R" phasecomponent signal of the receiver signal 37 can be detected with a firstphase reference signal on lead 42, and the "X" phase component signalcan be subsequently detected with a second phase reference signal onlead 42 that is orthogonal (90°) to the first.

The phase difference stored in waveform generator 26 is furtherillustrated as a function of time in FIG. 10. FIG. 10 shows a timingdiagram for the reference clock signal on lead 50 and the phasereference signal on lead 42 drawn relative to the transmitter signalvoltage waveform that is applied across the transmitter coil 34. For thepresently preferred embodiment, one cycle of the 10 KHz transmit signalfrequency is generated in 256 increments (this of course depends uponthe frequency selected, i.e., 128 for 20 KHz, 64 for 40 KHz).

Referring now to FIG. 10, the reference clock signal on lead 50 that isoutputted by the digital waveform generator 26 is shown phase shifted byan angle of θ₃. The phase reference signal on lead 42 is shown for boththe R and X phase relationships with the transmitter signal. Both the Rand X phase relationship waveforms for the phase reference signal areshown drawn as two separate signals, but in fact, there is only onephase reference signal 42 having one or the other phase relationshipillustrated depending on which of the quadrature phase components is tobe detected in the phase sensing means 43. FIG. 10 further illustratesthat for the phase reference signal, the total amount of phase shift isequal to the sum of the phase angles θ₃ +θ₄ (see also FIG. 4). The phaseangle θ₄ is that phase shift introduced in the phase reference signal onlead 42 by the autophase unit 32 while compensating for the dynamicphase shift errors.

The Autophase Unit 32

Referring now to FIG. 11, a detailed circuit diagram of the autophaseunit 32, the reference clock signal on lead 50 is shown applied to anR-2R precision resistor network 96. Responsive to a successiveapproximation register clock (SAR CLOCK) generated by the waveformgenerator 26 is a successive approximation register 90. The output fromthis register is also applied to the precision resistor network 96 sothat the voltage which appears on output line 108 of resistor network 96has a DC average value determined by the current digital content of thesuccessive approximation register 90. In this manner, the average valueof a filtered reference clock signal can be affected by controlling thedigital count in the successive approximation register 90.

Connectable from the output of the resistor network 96 to ground is aparallel combination of capacitors C4-C7, which are selectivelyconnected to ground through quad switch 100 in response to the frequencyselect control bits F₀ and F₁ inputted to the dual 2:4 line decoder 98.Decoder 98, switch 100 and capacitors C4-C7 comprise phase control means106 for controlling the phase shift applied to the filtered referredclock signal according to the magnitude of the contents of register 90.

The presence of a capacitor on the output of the resistor network 96functions as a low pass filter to filter or smooth the digital squarewave reference clock signal on lead 50 whose average value is beingmodulated by the value contained in the successive approximationregister 90. This voltage waveform 108 is illustrated in FIG. 10 showingthe smoothing effect of the filter capacitor connected to the output ofthe resistor network 96. Inverters 110 and 112, serially connected tothe output of the resistor network 96 function to convert the filteredand DC shifted reference clock signal into a square wave suitable fordigital circuits. A property of an inverter gate, such as inverter 110,is that the input DC voltage must exceed a threshold level before theoutput changes state. Thus, by smoothing the reference signal,controlling its DC average value and taking advantage of the thresholdproperty of an inverter gate, it is possible to affect a small amount ofphase shift to the reference signal as it appears on the output ofinverters 110 and 112. A phase shifted reference clock signal isillustrated in FIG. 10 as having been phase shifted in the amount of thephase angle θ₄.

Responding to the phase shifted reference clock signal on lead 113 arefirst and second flip-flops 102 and 104, respectively. These twoflip-flops are interconnected such that the output of flip-flop 102generates the phase reference signal on lead 42 which has either the "R"phase relationship or the "X" phase relationship to the transmitter coilcurrent i_(T) depending upon the phase select bits that are applied tothe decoder 98. These phase select bits are used to control the "preset"and "clear" inputs to the flip-flops.

By controlling the starting logic states of flip-flops 102 and 104 viathe preset and clear inputs, a square wave signal is generated on theoutput of flip-flop 102 that shift in phase by a precise 90° between thetwo phase relationships with respect to the transmitter current i_(T).This precise 90° phase changing relationship enables the phasequadrature component signals, "R" and "X", to be accurately detected bya single phase sensitive detector 40. Depending upon which of thetransmitter frequencies are selected, the quad switch 100 willselectively connect to the output of the resistor network 96 the properroll-off capacitance (C4-C7) to ground.

The Autocalibration Circuits

A common problem in signal processing circuits of the type adapted tothe borehole environment is a non-linearity of the transfer functionincluding non-linearities due to such things as temperature drift of thecomponents in the circuits. Also, the gain of devices may varysignificantly with signal level, and this problem is more apparent ininduction logging where signal levels vary over a wide range, forexample, several decades of magnitude. A single-point calibration may beentirely inadequate to characterize the system gain at a substantiallydifferent signal level. In particular, a system whose final result iscalculated as a small difference between two relatively large measuredsignals shows a particular sensitivity to small errors present as aresult of inaccurate calibration.

To substantially reduce this problem, the present invention includescalibration circuits that may be switched into the measurement channelat regular times intervals following a pre-programmed pattern. A sampleof the transmitter output is taken and used to develop calibrationsignals. FIG. 2 contains an illustration of the autocalibration circuitsaccording to the preferred embodiment of the invention to derivecalibration signals at the output tap points a, b, c and d oftransformer 29. Transformer 29 is designed to provide multiple signalsat levels sufficient to substantially cover the expected range offormation response signals. Controller 22 switches the measurementchannel sequentially between one of the calibration signals and thenormal receiver signals V_(r). During the autophase cycle previouslydiscussed, controllable switch S2 is selecting tap a as the test signalto be used. During an autocalibration cycle, anyone of the calibrationtaps may be used. Any changes in the output of the transmitter ormeasurement channel caused by temperature variations or other effects,will cause proportionate changes in both the calibration and receivedsignals, so that a compensation of this drift is possible.

A linearization formula can be obtained which takes the ratio of the twosignals and determines the characteristics of the formation relative tothe fixed values of the calibration circuitry. A mathematical procedureis used in CPU 10 to enable it to calculate the formula based on theresults of the calibration measurements representing the corrections tobe applied to subsequent formation signal data. This mathematicalprocedure in essence is a fitting of the calibration test signalresponse to a known mathematical function. This mathematical functionrepresents the calculated transfer function for the tool as obtainedfrom models of the circuit design of the tool using conventional circuitanalysis techniques. In effect, the test calibration points are used torecalculate the numerical constants in the transfer function.

These numerical constants used in the formula are periodicallyrecomputed at intervals of time determined by the anticipated magnitudeof temperature drift. The result of this process is shown in FIG. 13,where the transfer function of the tool is shown as a graph of outputsignals (S_(o)) against input signals (S_(i)). Four calibration points(X₁ -X₄) are shown, producing four outputs (Y₁ -Y₄). These outputs areused to compute a correction formula based on the knowledge of theamplitudes of X₁ -X₄. Subsequent formation data (S_(o)) applied by thecomputer to this formula will be substantially corrected to produce theapparent signal (S_(i)) in close conformity with S_(i). An analysis ofthe physical causes of non-linearity in the measurement channel of thetool allows the derivation of an equation that models the actualtransfer function. It is then possible to invert the equation to producethe correction formula. An alternative method, the use of a least meansquares fit of the formula, may also be used.

The multiple calibration points X₁ -X_(n) must satisfy two requirements;a precise ratio between each signal, and a constant absolute level ofthe ensemble. In the present invention (see FIG. 2), a precisionresistor R defines the constant absolute magnitude with transformer 29having multiple-ratio windings to supply the various temperature-stablecalibration signals. Thus, the requirement for hightemperature-stability is transferred from the measurement channelcircuits to the calibration circuits.

Noise signals may be unavoidably added to the formation signal either inthe formation or in the circuits of the tool. In both cases, the timespent by the system during calibration represents a loss of timeavailable to improve the signal-to-noise ratio by averaging (forexample, using CPU 10). Noise added in the tool introduces the extraburden of a reduction in calibration signal-to-noise ratio. The finalcomputed result S_(i) is a combination of formation and calibrationsignals, including their respective noise contributions. Since the twotypes of signals are measured sequentially, there will be no correlationbetween the noise components caused by random processes (for example,Johnson noise). An optimum solution must therefore be found for therelative times spent by the tool in measuring the two signals tomaximize the final signal-to-noise ratio.

To simplify the calculations, consider the case of a single calibrationpoint. This is an acceptable approximation if the amount ofnon-linearity in the system (and associated correction) is small. For asingle calibration point, the following obtains: ##EQU1## where S_(c)=calibration signal

S_(i) =formation signal

N_(i) =formation noise/unit bandwidth

N_(R) =receiver noise/unit bandwidth

T_(c) =calibration time

T_(m) =measurement time,

all referred, for example, to the receiver input.

The maximum time allowed to accurately measure S_(c) is determined bythe relatively slow drift of this parameter with changes in temperature,and for optimum S/N ratio the bandwidth should be no larger than thisvalue in the case of "white" noise with constant power per unitbandwidth. S_(i) must be measured with a bandwidth sufficient to resolvethe changing details of the rock formation as the logging tool traversesthe borehole. This will depend on the spatial resolution of thetransducer array and the speed at which the tool is moved. In bothcases, the bandwidths may be adjusted by the use of computer averagingof successive measurements. Where the logging speed is changed to suitdiffering operational requirements, the computer is programmed to adaptthe bandwidth automatically to optimize S/N ratio by averaging samplesas a function of distance along the borehole. When implemented, such asystem also allows the S/N ratio to be varied at will, by changing thelogging speed.

The final computation of the formation parameter (P) is done by takingthe ratio of the total formation signal to the total calibration signal.Since the noise sources are additive, we may use the method of partialderivatives to combine the separate signal/noise ratios: ##EQU2## Letthe total time available for a complete cycle of measurement andcalibration be,

    T=T.sub.m +T.sub.c,                                        (4)

then, ##EQU3## This function has a well defined maximum point whichoccurs at a measurement time fraction. ##EQU4## This analysis may beused to optimize the program of sequential measurements in the loggingtool. However, as already discussed, the final integration times (T_(m)and T_(c)) may be adjusted further by the computer to adapt toparticular situations.

The most efficient method of improving signal/noise ratio is for thetool to integrate the signal during successive time intervals devoted toeach of the sequential measurements. Floating point analog-to-digitalconverter 46 performs this function. The process is simplified if thesuccessive intervals are of identical length, but this may conflict withthe optimum formation/calibration time ratio calculated from equations(6) and (7).

A solution of the problem may be found by considering the pattern of thesamples. For example, where (T_(m) /T) optimum is approximately 0.75 or1/4, then we may make three formation measurements for everycalibration, i.e., calibration (tap "a"), formation, formation,formation, calibration (tap "b"), formation, etc. This pattern is storedin the controller 22, and may be modified to suit actual operatingconditions or changes in the performance of the tool.

The Floating Point Analog-to-Digital Converter 46

Turning now to FIG. 12, a detailed circuit diagram of a bipolar floatingpoint analog-to-digital converter is shown. The converter shown in FIG.12 may also operate as a unipolar converter, and the followingdescription refers to such operation. The output from the phasesensitive detector 40 is applied to a voltage-to-frequency converter 45which produces a digital clocking frequency signal 131 proportional tothe magnitude of the input analog voltage on lead 41. The basicprinciple of the converter of the present invention is to integrate thedetected component signal on lead 41 over a predetermined time intervalT_(i) by accumulating the clock cycles from the voltage-to-frequencyconverter 45 in two serially connected asynchronous 12-bit binarycounters 126 and 130. At the completion of each converter time intervalT_(i), the contents of counter 126,130 are transferred to a shiftregister consisting of serially connected shift register units 128, 132and 134.

Characteristic of digital induction logging tools is the very widedynamic range in signals that are generated on the output of phasesensitive detectors, such as the phase sensitive detector 40, as afunction of the conductivity of the formations being logged. In order tofunction properly over this wide dynamic range, a large number of binarycounter stages are required to accumulate the clock cycles from thevoltage-to-frequency converter 45 during each T_(i) interval. Eventhough a large number of clock cycles may be created, it is notnecessary that the full contents of the two binary counters 126, 130 betransmitted to the surface. Rather, data compression is used whereby thebinary number contained in counters 126 and 130 at the end of eachconversion interval are converted to a floating point number having adigital code for the magnitude and and a digital code for the exponent.The total number of digits for both the magnitude and the exponent isless than the total number of bits in counters 126 and 130. In thismanner, the digital samples obtained by the induction tool of thepresent invention are ready for floating point calculation by the CPU 10after transmission to the surface, and a significant reduction in theamount of data reported uphole is achieved.

In order to convert the accumulated conversion count in counter 126,130,the shift register 128,132,134 is loaded with the contents of the binarycounter 126,130 at the end of each conversion interval and shifted in adirection to cause the most significant bit (MSB) of a digital wordformed from a sub-set number of output bits of the shift register128,132,134 to contain a logic 1 (a logic 1 or logic 0 depending uponthe sign of the digital sample if bipolar operation is used). Thissub-set of output bits is less than the total number of bits containedin the binary counter 126,130, and represents the magnitude of thefloating point number thus obtained. As illustrated in FIG. 12, theoutput of the twelve most significant bits of the binary counter 126,130are presented as the magnitude of the floating point number. A downcounter 138 responds to the shift clock pulses that are shifting theshift register 128,130,134 so that for each shift pulse, down counter138 decrements its count by one count. The 4-bit down counter 138comprises the 4-bit exponent of the floating point conversion value. Atthe start of each floating point conversion, the contents of downcounter 138 is preset to an all 1's pattern.

The shifting of the contents contained in shift register 128,132,134 iscontinually shifted to the left until one of two events occurs, either alogic 1 appears in the most significant bit of the magnitude code wordfor the floating point value or the count in down counter 138 reacheszero. Upon the occurrence of either of these events, shifting of shiftregister 128,132,134 ceases, and both the magnitude and the exponentvalues are transmitted as the floating point conversion sample from A/Dconverter 46. In a further implementation, the most significant bit(MSB) containing a logic 1 or logic 0 as previously described iseliminated by shifting the data one additional time, and means areprovided in subsequent decoding to restore the missing bit, allowing anincrease in the accuracy of the floating point sample.

The length of the conversion cycle time T_(i) and the constant ofproportionality for the voltage-to-frequency converter 45 determines theconversion constant for converting the digital magnitude and exponentfloating point values to an amplitude value for the detected phasecomponent signal on lead 41. For unipolar operation,voltage-to-frequency converter 45 has a conversion coefficient ofbetween 200 and 500 KHz/volt and a dynamic range of 1 millivolt to 10volts (80 db) at all temperatures from -65° C. to 200° C.

Each of the circuits shown and described herein are intended to operatein extreme temperature environments encountered in induction welllogging. Temperatures in excess of 200° C. are not uncommon.Accordingly, where such temperatures are to be encountered, attentionmust be given to circuit component selection, layout and design in orderto insure that the circuits will continue to perform their functions asdisclosed and discussed herein.

In FIG. 6 is illustrated a timing diagram for various of the signalspresent in the bipolar A/D converter circuit shown in FIG. 12. Theoperation of converter 46 when operating in a bipolar mode isessentially the same as described above with reference to operation in aunipolar mode except that counter 126 and 130 are disposed to count upor down depending on the polarity of the input. In bipolar operation,the input voltage 41 from the phase sensitive detector 40 will havevoltage excursions both positive and negative. In unipolar operation,the voltage excursions are all positive. FIG. 6 illustrates a possiblebipolar voltage signal for the signal on lead 41. To operate as abipolar A/D converter, the voltage-to-frequency converter 45 describedabove must operate to produce an output frequency signal whose frequencyis dependent upon the absolute value of the magnitude of its inputsignal. That is, produce the same frequency for a given positive voltageas it does for a negative voltage of the same magnitude.

To distinguish between positive and negative input voltage, a comparator127 responds to the signal on lead 41 to produce a counting controlsignal, UP/DOWN CONTROL, to up/down counters 126 and 130. As shown inFIG. 6, when the signal on lead 41 is positive, counter 126 and 130count up, and when negative, the counters count down. In this manner,the count in counter 126 and 130 at the end of each counting periodT_(i) contains the average value of input signal 41 over one timeperiod, and the most significant bit in the last counter of the chain incounters 126 and 130 represents the polarity of the average value.

At the completion of each counting interval T_(i), RESET clears counter126 and 130, and the next counting interval is begun. At the same time,down counter 138 is set to the all 1's pattern so that the count for thejust completed counting interval, which is now contained in shiftregisters 128, 132, 134, can be converted to a floating point sample aspreviously described. In other words, as each counting interval isbegun, the previous integrated sample obtained by converter 46 isconverted to a floating point sample and made available for transmissionto the surface.

Summary of Operation

In normal operation, the digital induction logging tool 1 of the presentinvention responds to command and data signals transmitted from asurface located central processing unit 10 to select a transmitterfrequency from among a plurality of transmitter frequencies or a choiceof multiple simultaneous frequencies. The frequency or frequenciesselected may be based on such things as the actual conductivity of theformation being measured in order to automatically obtain the mostaccurate reading of conductivity possible. A digital waveform generator26 responds to the frequency select signals to digitally generate a lowdistortion, highly phase stable sinusoidal transmitter frequency signalor a waveform consisting of two or more superimposed sinusoidal signals.This transmitter signal is applied to a transmitter coil 34. Using theprinciple of superposition we may consider the different frequencies ofa composite waveform as described above as if they were separate. Thefollowing discussion will consider the case of a sinusoidal transmitteroutput.

Responsive to the magnetic field generated in the earth's formations bythe current flowing in this transmitter coil, formation eddy currentsare caused to flow. These flowing eddy currents themselves producemagnetic fields which are sensed by a receiver coil 36 to produce areceive signal V_(r) on lead 37. A single phase sensitive detector 40 isemployed to sequentially detect, at each measurement point of theborehole the phase quadrature components, "R" and "X", contained in thereceiver signal to obtain signals that are representative of theconductivity of the formations.

Although the R and X components are sequentially measured in a singlephase sensitive detector by the present invention, the rate at whichthese measurements are sequentially taken is high enough that both the Rand X measurements for a given depth point are the same as those thatwould have been obtained had the tool been stationary while the R and Xmeasurements were made. In other words, the present invention obtainsessentially correlated R and X readings for each depth point even thoughthe measurements are sequentially made. In order to achieve precise andaccurate measurements of these phase quadrature component signals, thepresent invention compensates for phase shift errors resulting fromcomponents of the circuits in the induction tool itself. These phaseshift errors are characterized in two classifications, static anddynamic or temperature dependent varying phase shift errors.

The digital waveform generator 26 also produces a reference clock signalon lead 50 having a predetermined phase relationship to the resultingsinusoidal transmitter signal on lead 28, which the waveform generatoralso produces. Responsive to the reference clock signal, an autophaseunit 32 generates a phase reference signal on lead 42 to a phasesensitive detector 40. The phase shift between the reference clock andthe transmitter signal compensates for the static phase shift errors. Tocompensate for the temperature dependent phase shift errors, anautophase cycle is periodically initiated during a normal logging run.During this autophase cycle, a predetermined test signal derived fromthe transmitter current i_(T) is to the detector 40, rather than thenormal receiver signal V_(r) from the receiver coil 36. Additionally,the phase relationship of the phase reference signal on lead 42 isselected to detect the "X" phase component signal.

The polarity of the resulting output of the phase sensitive detector 40is detected and applied as a feedback error signal to the autophase unit32. This feedback signal controls the autophase unit 32 to phase shiftthe phase reference signal on lead 42 in a direction to reduce theoutput of the phase sensitive detector 40 to zero. Because there shouldbe zero "X" component signal in the test signal derived from thetransmitter current i_(T), it follows that by phase shifting the phasereference signal in a direction to insure that the detected phasecomponent signal is actually zero automatically and dynamicallycompensates for any temperature dependent phase shift errors present inall the measurement of either the R or the X phase component signals.

During the normal logging run, the R and X phase component signalssequentially detected by the single phase sensitive detector 40 areconverted to floating point digital samples by a wide dynamic rangefloating point converter 46. Thus, the present invention is able toobtain very precise and accurate digital measurements of the R and the Xcomponent signals in the receiver signal V_(r) downhole where thesignal-to-noise ratio is at its maximum before being transmitted to thesurface CPU 10 for further processing.

During the sequential measurement of components in the receiver signal,the tool produces a sequence of predetermined calibration signal derivedfrom the transmitter current i_(T) which are applied to the phasesensitive detector 40 in place of the normally applied receiver signalV_(r). From the measurements thus obtained for the calibration signals,the surface located CPU 10 is able to produce a correction formula forcorrecting the signal measurements to eliminate the non-linearities inthe transfer functions of the tool itself. These non-linearities resultfrom such things as non-linearities in gain over wide signal ranges andin temperature drift in the circuit components.

In describing the invention, reference has been made to a preferredembodiment. However, those skilled in the art and familiar with thedisclosure of the invention may recognize additions, deletions,substitutions or other modifications, which would fall within thepurview of the invention as defined in the appended claims.

I claim:
 1. An induction logging tool adapted for operation in aborehole in association with surface equipment, for measuring acharacteristic of sub-surface formations, the tool including atransmitter coil which induces formation currents to flow in response toa transmitter signal and a receiver coil which generates a signalcharacteristic of the formation in response to these currents, the toolcomprising:a waveform generator for digitally generating a sinusoidaltransmitter signal and a reference clock signal, said generatorincluding:a digital memory having a plurality of address sections, saidaddress sections having stored therein magnitude values of sinusoidalwaveforms of a respective plurality of different predeterminedfrequencies, said memory having further stored therein reference clocksignal generating data, said data being stored in memory addresslocations selected relative to said magnitude values in such manner thatthe reference clock signal generated from said data is phase shifted apredetermined amount with respect to the transmitter signal of therespective frequency, said memory having a frequency select addressportion the content of which specifies a particular address section,latch means providing the content of said frequency select addressportion in response to commands received from the surface equipment, aphase shift unit responsive to said reference clock signal forgenerating a reference phase signal, and a phase sensitive detectorresponsive to said reference phase signal and said characteristic signalfrom the receiver coil for generating an output signal indicative of themagnitude of a component of the characteristic signal that is in-phasewith the reference phase signal.